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pin connections 8-lead narrow-body so 8-lead epoxy dip (s suffix) (p suffix) out a ?n a +in a v v+ out b ?n b +in b 1 2 3 4 5 6 7 8 op295 out a ?n a +in a v v+ out b ?n b +in b 1 2 3 45 6 7 8 op295 14-lead epoxy dip 16-lead so (300 mil) (p suffix) (s suffix) out a ?n a +in a v out b ?n d +in d out d ?n b +in b v+ out c ?n c +in c 1 2 3 4 11 12 13 14 op495 5 6 7 8 9 10 out a ?n a +in a v out d ?n d +in d out b ?n b +in b nc v+ out c ?n c +in c 1 2 3 413 14 15 16 5 6 7 89 10 11 12 op495 nc nc = no connect rev. b information furnished by analog devices is believed to be accurate and reliable. however, no responsibility is assumed by analog devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. no license is granted by implication or otherwise under any patent or patent rights of analog devices. a dual/quad rail-to-rail operational amplifiers op295/op495 ? analog devices, inc., 1995 one technology way, p.o. box 9106, norwood. ma 02062-9106, u.s.a. tel: 617/329-4700 fax: 617/326-8703 features rail-to-rail output swing single-supply operation: +3 v to +36 v low offset voltage: 300 m v gain bandwidth product: 75 khz high open-loop gain: 1000 v/mv unity-gain stable low supply current/per amplifier: 150 m a max applications battery operated instrumentation servo amplifiers actuator drives sensor conditioners power supply control general description rail-to-rail output swing combined with dc accuracy are the key features of the op495 quad and op295 dual cbcmos opera- tional amplifiers. by using a bipolar front end, lower noise and higher accuracy than that of cmos designs has been achieved. both input and output ranges include the negative supply, pro- viding the user zero-in/zero-out capability. for users of 3.3 volt systems such as lithium batteries, the op295/op495 is specified for three volt operation. maximum offset voltage is specified at 300 m v for +5 volt opera- tion, and the open-loop gain is a minimum of 1000 v/mv. this yields performance that can be used to implement high accuracy systems, even in single supply designs. the ability to swing rail-to-rail and supply +15 ma to the load makes the op295/op495 an ideal driver for power transistors and h bridges. this allows designs to achieve higher efficien- cies and to transfer more power to the load than previously pos- sible without the use of discrete components. for applications that require driving inductive loads, such as transformers, in- creases in efficiency are also possible. stability while driving capacitive loads is another benefit of this design over cmos rail-to-rail amplifiers. this is useful for driving coax cable or large fet transistors. the op295/op495 is stable with loads in excess of 300 pf. the op295 and op495 are specified over the extended indus- trial (C40 c to +125 c) temperature range. op295s are avail- able in 8-pin plastic and ceramic dip plus so-8 surface mount packages. op495s are available in 14-pin plastic and so-16 surface mount packages. contact your local sales office for mil-std-883 data sheet.
rev. b C2C op295/op495Cspecifications electrical characteristics parameter symbol conditions min typ max units input characteristics offset voltage v os 30 300 m v C40 c t a +125 c 800 m v input bias current i b 820 na C40 c t a +125 c30na input offset current i os 1 3na C40 c t a +125 c 5na input voltage range v cm 0 +4.0 v common-mode rejection ratio cmrr 0 v v cm 4.0 v, C40 c t a +125 c 90 110 db large signal voltage gain a vo r l = 10 k w , 0.005 v out 4.0 v 1000 10,000 v/mv r l = 10 k w , C40 c t a +125 c 500 v/mv offset voltage drift d v os / d t 15 m v/ c output characteristics output voltage swing high v oh r l = 100 k w to gnd 4.98 5.0 v r l = 10 k w to gnd 4.90 4.94 v i out = 1 ma, C40 c t a +125 c 4.7 v output voltage swing low v ol r l = 100 k w to gnd 0.7 2 mv r l = 10 k w to gnd 0.7 2 mv i out = 1 ma, C40 c t a +125 c90mv output current i out 11 18 ma power supply power supply rejection ratio psrr 1.5 v v s 15 v 90 110 db 1.5 v v s 15 v, C40 c t a +125 c85db supply current per amplifier i sy v out = 2.5 v, r l = , C40 c t a +125 c 150 m a dynamic performance skew rate sr r l = 10 k w 0.03 v/ m s gain bandwidth product gbp 75 khz phase margin q o 86 degrees noise performance voltage noise e n p-p 0.1 hz to 10 hz 1.5 m v p-p voltage noise density e n f = 1 khz 51 nv/ ? hz current noise density i n f = 1 khz <0.1 pa/ ? hz specifications subject to change without notice. electrical characteristics parameter symbol conditions min typ max units input characteristics offset voltage v os 30 500 m v input bias current i b 820 na input offset current i os 1 3na input voltage range v cm 0 +2.0 v common-mode rejection ratio cmrr 0 v v cm 2.0 v, C40 c t a +125 c 90 110 db large voltage gain a vo r l = 10 k w 750 v/mv offset voltage drift d v os / d t 1 m v/ c output characteristics output voltage swing high v oh r l = 10 k w to gnd 2.9 v output voltage swing low v ol r l = 10 k w to gnd 0.7 2 mv power supply power supply rejection ratio psrr 1.5 v v s 15 v 90 110 db 1.5 v v s 15 v, C40 c t a +125 c85db supply current per amplifier i sy v out = 1.5 v, r l = , C40 c t a +125 c 150 m a dynamic performance slew rate sr r l = 10 k w 0.03 v/ m s gain bandwidth product gbp 75 khz phase margin q o 85 degrees noise performance voltage noise e n p-p 0.1 hz to 10 hz 1.6 m v p-p voltage noise density e n f = 1 khz 53 nv/ ? hz current noise density i n f = 1 khz <0.1 pa/ ? hz specifications subject to change without notice. (@ v s = +5.0 v, v cm = +2.5 v, t a = +25 8 c unless otherwise noted) (@ v s = +3.0 v, v cm = +1.5 v, t a = +25 8 c unless otherwise noted)
electrical characteristics parameter symbol conditions min typ max units input characteristics offset voltage v os 30 300 m v C40 c t a +125 c 800 m v input bias current i b v cm = 0 v 7 20 na v cm = 0 v, C40 c t a +125 c30na input offset current i os v cm = 0 v 1 3na v cm = 0 v, C40 c t a +125 c 5na input voltage range v cm C15 +13.5 v common-mode rejection ratio cmrr C 15.0 v v cm +13.5 v, C40 c t a +125 c 90 110 db large signal voltage gain a vo r l = 10 k w 1000 4000 v/mv offset voltage drift d v os / d t 1 m v/ c output characteristics output voltage swing high v oh r l = 100 k w to gnd 14.95 v r l = 10 k w to gnd 14.80 v output voltage swing low v ol r l = 100 k w to gnd C14.95 v r l = 10 k w to gnd C14.85 v output current i out 15 25 ma power supply power supply rejection ratio psrr v s = 1.5 v to 15 v 90 110 db v s = 1.5 v to 15 v, C40 c t a +125 c85 db supply current i sy v o = 0 v, r l = , v s = 18 v, C40 c t a +125 c 175 m a supply voltage range v s +3 ( 1.5) +36 ( 18) v dynamic performance slew rate sr r l = 10 k w 0.03 v/ m s gain bandwidth product gbp 85 khz phase margin q o 83 degrees noise performance voltage noise e n p-p 0.1 hz to 10 hz 1.25 m v p-p voltage noise density e n f =1 khz 45 nv/ ? hz current noise density i n f = 1 khz <0.1 pa/ ? hz specifications subject to change without notice. wafer test limits parameter symbol conditions limit units offset voltage vos 300 m v max input bias current i b 20 na max input offset current i os 2 na max input voltage range 1 v cm 0 to +4 v min common-mode rejection ratio cmrr 0 v v cm 4 v 90 db min power supply rejection ratio psrr 1.5 v v s 15 v 90 m v/v large signal voltage gain a vo r l = 10 k w 1000 v/mv min output voltage swing high v oh r l = 10 k w 4.9 v min supply current per amplifier i sy v out = 2.5 v, r l = 150 m a max notes electrical tests and wafer probe to the limits shown. due to variations in assembly methods and normal yield loss, yield after packaging is not guaranteed for standard product dice. consult factory to negotiate specifications based on dice lot qualifications through sample lot assembly and testing. 1 guaranteed by cmrr test. ordering guide op295/op495 rev. b C3C (@ v s = 15.0 v, t a = +25 8 c unless otherwise noted) temperature package package model range description option op295gp C40 c to +125 c 8-pin plastic dip n-8 op295gs C40 c to +125 c 8-pin soic so-8 OP295GBC +25 c dice temperature package package model range description option op495gp C40 c to +125 c 14-pin plastic dip n-14 op495gs C40 c to +125 c 16-pin sol r-16 op495gbc +25 c dice (@ v s = +5.0 v, v cm = 2.5 v, t a = +25 8 c unless otherwise noted)
rev. b C4C op295/op495 absolute maximum ratings 1 supply voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 v input voltage 2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 v differential input voltage 2 . . . . . . . . . . . . . . . . . . . . . . . +36 v output short-circuit duration . . . . . . . . . . . . . . . . . indefinite storage temperature range p, s package . . . . . . . . . . . . . . . . . . . . . . . . C65 c to +150 c operating temperature range op295g, op495g . . . . . . . . . . . . . . . . . . . C40 c to +125 c junction temperature range p, s package . . . . . . . . . . . . . . . . . . . . . . . . C65 c to +150 c lead temperature range (soldering, 60 sec) . . . . . . . +300 c package type u ja 3 u jc unit 8-pin plastic dip (p) 103 43 c/w 8-pin soic (s) 158 43 c/w 14-pin plastic dip (p) 83 39 c/w 16-pin so (s) 98 30 c/w notes 1 absolute maximum ratings apply to both dice and packaged parts, unless otherwise noted. 2 for supply voltages less than 18 v, the absolute maximum input voltage is equal to the supply voltage. 3 q ja is specified for the worst case conditions, i.e., q ja is specified for device in socket for cerdip, p-dip, and lcc packages; q ja is specified for device soldered in circuit board for soic package. dice characteristics op295 die size 0.066 0.080 inch, 5,280 sq. mils. substrate (die backside) is connected to v+. transistor count, 74. op495 die size 0.113 0.083 inch, 9,380 sq. mils. substrate (die backside) is connected to v+. transistor count, 196. typical characteristics 140 20 100 80 40 ?5 60 ?0 120 100 75 50 25 0 temperature ? c supply current per amplifier ? m a v s = +5v v s = +3v v s = +36v 15.2 ?5.2 100 ?4.6 ?5.0 ?5 ?4.8 ?0 14.2 ?4.4 14.4 14.6 14.8 15.0 75 50 25 0 temperature ? c ?output swing ?volts + output swing ?volts v s = 15v r l = 100k r l = 2k r l = 2k r l = 100k r l = 10k r l = 10k supply current per amplifier vs. temperature output voltage swing vs. temperature
rev. b C5C typical characteristicsCop295/op495 3.10 2.50 100 2.80 2.60 ?5 2.70 ?0 3.00 2.90 75 50 25 0 temperature ? c output voltage swing ?volts r l = 2k v s = +3v r l = 100k r l = 10k output voltage swing vs. temperature 200 0 250 50 25 ?00 ?50 100 75 125 150 175 200 150 100 50 0 ?0 ?00 ?50 v s = +5v t a = +25 c input offset voltage ? m v units based on 600 op amps op295 input offset (v os ) distribution units 250 0 3.2 75 25 0.4 50 0 150 100 125 175 200 225 2.8 2.4 2.0 1.6 1.2 0.8 based on 600 op amps v s = +5v ?0 t a +85 c t c ?v os ? m v/ c op295 tcCv os distribution 5.10 4.50 100 4.80 4.60 ?5 4.70 ?0 5.00 4.90 75 50 25 0 temperature ? c output voltage swing ?volts v s = +5v r l = 100k r l = 2k r l = 10k output voltage swing vs. temperature 500 0 300 150 50 ?0 100 ?00 300 200 250 350 400 450 250 200 150 100 50 0 input offset voltage ? m v units v s = +5v t a = +25 c based on 1200 op amps op495 input offset (v os ) distribution 500 0 3.2 150 50 0.4 100 0 300 200 250 350 400 450 2.8 2.4 2.0 1.6 1.2 0.8 t c ?v os ? m v/ c units v s = +5v ?0 t a +85 c based on 1200 op amps op495 tcCv os distribution
rev. b C6C op295/op495Ctypical characteristics v s = +5v 20 0 100 12 4 ?5 8 ?0 16 75 50 25 0 temperature ? c input bias current ?na input bias current vs. temperature v s = +5v temperature ? c 40 0 100 10 5 ?5 ?0 20 15 25 30 35 75 50 25 0 v s = 15v source sink source sink output current ?ma output current vs. temperature temperature ? c open-loop gain ?v/ m v r l = 2k 100 10 1 ?0 25 100 0 ?5 50 75 v s = 15v v o = 10v r l = 10k r l = 100k open-loop gain vs. temperature 12 0 100 6 2 ?5 4 ?0 10 8 75 50 25 0 temperature ? c open-loop gain ?v/ m v v s = +5v v o = +4v r l = 100k r l = 10k r l = 2k open-loop gain vs. temperature 1v 100 m v 1 m a10 m a 10ma 1ma 100 m a 100mv 10mv 1mv load current output voltage d to rail source sink v s = +5v t a = +25 c output voltage to supply rail vs. load current
op295/op495 rev. b C7C applications rail-to-rail applications information the op295/op495 has a wide common-mode input range ex- tending from ground to within about 800 mv of the positive supply. there is a tendency to use the op295/op495 in buffer applications where the input voltage could exceed the common- mode input range. this may initially appear to work because of the high input range and rail-to-rail output range. but above the common-mode input range the amplifier is, of course, highly nonlinear. for this reason it is always required that there be some minimal amount of gain when rail-to-rail output swing is desired. based on the input common-mode range this gain should be at least 1.2. low drop-out reference the op295/op495 can be used to gain up a 2.5 v or other low voltage reference to 4.5 volts for use with high resolution a/d converters that operate from +5 volt only supplies. the circuit in figure 1 will supply up to 10 ma. its no-load drop-out volt- age is only 20 mv. this circuit will supply over 3.5 ma with a +5 volt supply. 16k v out = 4.5v 1 to 10 m f 10 w 0.001 m f 20k ref43 2 6 4 +5v +5v 1/2 op295/ op495 figure 1. 4.5 volt, low drop-out reference low noise, single supply preamplifier most single supply op amps are designed to draw low supply current, at the expense of having higher voltage noise. this tradeoff may be necessary because the system must be powered by a battery. however, this condition is worsened because all circuit resistances tend to be higher; as a result, in addition to the op amps voltage noise, johnson noise (resistor thermal noise) is also a significant contributor to the total noise of the system. the choice of monolithic op amps that combine the characteris- tics of low noise and single supply operation is rather limited. most single supply op amps have noise on the order of 30 nv/ ? hz to 60 nv/ ? hz and single supply amplifiers with noise below 5 nv/ ? hz do not exist. in order to achieve both low noise and low supply voltage opera- tion, discrete designs may provide the best solution. the circuit on figure 2 uses the op295/op495 rail-to-rail amplifier and a matched pnp transistor pairthe mat03to achieve zero-in/ zero-out single supply operation with an input voltage noise of 3.1 nv/ ? hz at 100 hz. r5 and r6 set the gain of 1000, making this circuit ideal for maximizing dynamic range when amplifying low level signals in single supply applications. the op295/op495 provi des rail-to-rail output swings, allowing this circuit to oper- ate with 0 to 5 volt outputs. only half of the op295/op495 is used, leaving the other uncommitted op amp for use elsewhere. mat- 03 5 6 3 2 1 7 r3 r4 q1 q2 r1 r6 10 w v out led 1 8 4 3 2 r5 10k w c2 10 m f r7 51 0 w c1 1500pf r8 100 w r2 27k w v in 0.1 m f 10 m f q2 2n3906 op295/ op495 figure 2. low noise single supply preamplifier the input noise is controlled by the mat03 transistor pair and the collector current level. increasing the collector current re- duces the voltage noise. this particular circuit was tested with 1.85 ma and 0.5 ma of current. under these two cases, the in- put voltage noise was 3.1 nv/ ? hz and 10 nv/ ? hz , respectively. the high collector currents do lead to a tradeoff in supply cur- rent, bias current, and current noise. all of these parameters will increase with increasing collector current. for example, typically the mat03 has an h fe = 165. this leads to bias currents of 11 m a and 3 m a, respectively. based on the high bias currents, this circuit is best suited for applications with low source imped- ance such as magnetic pickups or low impedance strain gages. furthermore, a high source impedance will degrade the noise performance. for example, a 1 k w resistor generates 4 nv/ ? hz of broadband noise, which is already greater than the noise of the preamp. the collector current is set by r1 in combination with the led and q2. the led is a 1.6 v zener that has a temperature co- efficient close to that of q2s base-emitter junction, which pro- vides a constant 1.0 v drop across r1. with r1 equal to 270 w , the tail current is 3.7 ma and the collector current is half that, or 1.85 ma. the value of r1 can be altered to adjust the collec- tor current. whenever r1 is changed, r3 and r4 should also be adjusted. to maintain a common-mode input range that in- cludes ground, the collectors of the q1 and q2 should not go above 0.5 votherwise they could saturate. thus, r3 and r4 have to be small enough to prevent this condition. their values and the overall performance for two different values of r1 are summarized in table i. lastly, the potentiometer, r8, is needed to adjust the offset voltage to null it to zero. similar perfor- mance can be obtained using an op90 as the output amplifier with a savings of about 185 m a of supply current. however, the output swing will not include the positive rail, and the band- width will reduce to approximately 250 hz.
rev. b C8C op295/op495 table i. single supply low noise preamp performance i c = 1.85 ma i c = 0.5 ma r1 270 w 1.0 k w r3, r4 200 w 910 w e n @ 100 hz 3.15 nv/ ? hz 8.6 nv/ ? hz e n @ 10 hz 4.2 nv/ ? hz 10.2 nv/ ? hz i sy 4.0 ma 1.3 ma i b 11 m a3 m a bandwidth 1 khz 1 khz closed-loop gain 1000 1000 driving heavy loads the op295/op495 is well suited to drive loads by using a power transistor, darlington or fet to increase the current to the load. the ability to swing to either rail can assure that the device is turned on hard. this results in more power to the load and an increase in efficiency over using standard op amps with their limited output swing. driving power fets is also possible with the op295/op495 because of its ability to drive capacitive loads of several hundred picofarads without oscillating. without the addition of external transistors the op295/op495 can drive loads in excess of 15 ma with 15 or +30 volt supplies. this drive capability is somewhat decreased at lower supply voltages. at 5 volt supplies the drive current is 11 ma. driving motors or actuators in two directions in a single supply application is often accomplished using an h bridge. the principle is demonstrated in figure 3a. from a single +5 volt supply this driver is capable of driving loads from 0.8 v to 4.2 v in both directions. figure 3b shows the voltages at the inverting and noninverting outputs of the driver. there is a small crossover glitch that is frequency dependent and would not cause problems 5k 10k 1.67v 10k 10 k 2n2222 2n2222 outputs 2n2907 2n2907 +5 v 0 v in 2.5v figure 3a. h bridge 10 90 100 0% 2v 2v 1ms figure 3b. h bridge outputs unless this was a low distortion application such as audio. if this is used to drive inductive loads, be sure to add diode clamps to protect the bridge from inductive kickback. direct access arrangement op295/op495 can be used in a single supply direct access ar- rangement (daa) as is shown an in figure 4. this figure shows a portion of a typical dm capable of operating from a single +5 volt supply and it may also work on +3 volt supplies with minor modifications. amplifiers a2 and a3 are configured so that the transmit signal txa is inverted by a2 and is not in- verted by a3. this arrangement drives the transformer differen- tially so that the drive to the transformer is effectively doubled over a single amplifier arrangement. this application takes ad- vantage of the op295/op495s ability to drive capacitive loads, and to save power in single supply applications. 2.5v ref a3 20k w 20k w 750pf 20k w 22.1k w 0.1 m f 475 w 3.3k w 0.0047 m f 0.1 m f 0.033 m f a2 a1 20k w 20k w 37.4k w 390pf rxa txa op295/ op495 op295/ op495 op295/ op495 1:1 figure 4. direct access arrangement a single supply instrumentation amplifier the op295/op495 can be configured as a single supply instru- mentation amplifier as in figure 5. for our example, v ref is set equal to v + 2 and v o is measured with respect to v ref . the in- put common-mode voltage range includes ground and the out- put swings to both rails. v+ v in v ref 1 r1 100k r2 20k r3 20k r4 100k v o v in + v ref v o = 5 + 200k r g ( ) r g 3 2 8 7 4 5 6 1/2 op295/ op495 1/2 op295/ op495 figure 5. single supply instrumentation amplifier resistor r g sets the gain of the instrumentation amplifier. mini- mum gain is 6 (with no r g ). all resistors should be matched in absolute value as well as temperature coefficient to maximize
op295/op495 rev. b C9C common-mode rejection performance and minimize drift. this instrumentation amplifier can operate from a supply voltage as low as 3 volts. a single supply rtd thermometer amplifier this rtd amplifier takes advantage of the rail-to-rail swing of the op295/op495 to achieve a high bridge voltage in spite of a low 5 v supply. the op295/op495 amplifier servos a constant 200 m a current to the bridge. the return current drops across the parallel resistors 6.19 k w and the 2.55 m w , developing a voltage that is servoed to 1.235 v, which is established by the ad589 bandgap reference. the 3-wire rtd provides an equal line resistance drop in both 100 w legs of the bridge, thus im- proving the accuracy. the amp04 amplifies the differential bridge signal and converts it to a single-ended output. the gain is set by the series resis- tance of the 332 w resistor plus the 50 w potentiometer. the gain scales the output to produce a 4.5 v full scale. the 0.22 m f capacitor to the output provides a 7 hz low-pass filter to keep noise at a minimum. 3 7 1 8 6 5 4 2 amp04 0.22 m f 332 w 50 w 26.7k 0.5% 100 w 0.5% 100 w rtd 2.55m 1% 6.19k 1% ad589 1.235 37.4k +5v 200 w 10-turns zero adj 26.7k 0.5% v o 4.5v = 450 c 0v = 0 c +5v 1 2 3 1/2 op295/ op495 figure 6. low power rtd amplifier a cold junction compensated, battery powered thermocouple amplifier the op295/op495s 150 m a quiescent current per amplifier consumption makes it useful for battery powered temperature measuring instruments. the k-type thermocouple terminates into an isothermal block where the terminated junctions ambi- ent temperatures can be continuously monitored and corrected by summing an equal but opposite thermal emf to the ampli- fier, thereby canceling the error introduced by the cold ju nctions. v o 0v = 0 c 5v = 500 c 4.99k 1% 1.33m w 20k scale adjust 9v 24.3k 1% 24.9k 7.15k 1% 1.235v ad589 24.9k 1% 500 w 10-turn 2.1k 1% 475 w 1% 1.5m 1% 1n914 isothermal block cold junctions cr al chromel k-type thermocouple 40.7 m v/ c zero adjust 8 1 4 2 3 alumel op295/ op495 figure 7. battery powered, cold-junction compensated thermocouple amplifier to calibrate, immerse the thermocouple measuring junction in a 0 c ice bath, adjust the 500 w zero adjust pot to zero volts out. then immerse the thermocouple in a 250 c temperature bath or oven and adjust the scale adjust pot for an output voltage of 2.50 v, which is equivalent to 250 c. within this temperature range, the k-type thermocouple is quite accurate and produces a fairly linear transfer characteristic. accuracy of 3 c is achiev- able without linearization. even if the battery voltage is allowed to decay to as low as 7 volts, the r ail-to-rail swing allows temperature measurem ents to 700 c. however, linearization may be necessary for tempera- tures above 250 c where the thermocouple becomes rather nonlinear. the circuit draws just under 500 m a supply current from a 9 v battery. a 5 v only, 12-bit dac that swings 0 v to 4.095 v figure 8 shows a complete voltage output dac with wide out- put voltage swing operating off a single +5 v supply. the serial input 12-bit d/a converter is configured as a voltage output device with the 1.235 v reference feeding the current output pin (i out ) of the dac. the v ref which is normally the input now becomes the output. the output voltage from the dac is the binary weighted volt- age of the reference, which is gained up by the output amplifier such that the dac has a 1 mv per bit transfer function. +5v r2 41.2k r3 5k w r4 100k w v dd r fb v ref sri clk gnd i out dac8043 3 4 7 65 1 2 8 +5v digital control +5v ad589 r1 17.8k w +1.23v ld v o = (4.096v) d 4096 total power dissipation = 1.6mw 8 1 4 2 3 op295/ op495 figure 8. a 5 volt 12-bit dac with 0 v to +4.095 output swing 4C20 ma current loop transmitter figure 9 shows a self powered 4C20 ma current loop transmit- ter. the entire circuit floats up from the single supply (12 v to 36 v) return. the supply current carries the signal within the 4 to 20 ma range. thus the 4 ma establishes the baseline 8 1 2 3 220pf 220 w ref02 gnd 62 4 100 w 2n1711 100k 1% hp 5082-2800 100 w 1% v in 0 + 3v 100k w 10-turn 1.21m 1% null adj span adj 182k 1% 10k w 10-turn 5v +1 2v to +3 6v r l 100 w 4?0ma 4 1/2 op295/ op495 figure 9. 4C20 ma current loop transmitter
rev. b C10C op295/op495 current budget with which the circuit must operate. this circuit consumes only 1.4 ma maximum quiescent current, making 2.6 ma of current available to power additional signal conditioning circuitry or to power a bridge circuit. a 3 volt low-dropout linear voltage regulator figure 10 shows a simple 3 v voltage regulator design. the regulator can deliver 50 ma load current while allowing a 0.2 v dropout voltage. the op295/op495s rail-to-rail output swing handily drives the mje350 pass transistor without requiring spe- cial drive circuitry. at no load, its output can swing less than the pass transistors base-emitter voltage, turning the device nearly off. at full load, and at low emitter-collector voltages, the tran- sistor beta tends to decrease. the additional base current is eas- ily handled by the op295/op495 output. the amplifier servos the output to a constant voltage, which feeds a portion of the signal to the error amplifier. higher output current, to 100 ma, is achievable at a higher dropout voltage of 3.8 v. 1000pf 43k 44.2k 1% 30.9k 1% ad589 1.235v v o 100 m f i l < 50ma mje 350 v in 5v to 3.2v 1/2 op295/ op495 8 1 4 2 3 figure 10. 3 v low dropout voltage regulator figure 11 shows the regulators recovery characteristic when its output underwent a 20 ma to 50 ma step current change. 10 100 0% 90 1ms 20mv 2v 50ma 20ma output step current control waveform figure 11. output step load current recovery low-dropout, 500 ma voltage regulator with fold-back current limiting adding a second amplifier in the regulation loop as shown in figure 12 provides an output current monitor as well as fold- back current limiting protection. amplifier a1 provides error amplification for the normal voltage regulation loop. as long as the output current is less than 1 am- pere, amplifier a2s output swings to ground, reverse biasing the diode and effectively taking itself out of the circuit. however, as the output current exceeds 1 amp, the voltage that develops across the 0.1 w sense resistor forces the amplifier a2s output to go high, forward-biasing the diode, which in turn closes the current limit loop. at this point a2s lower output resistance dominates the drive to the power mosfet transistor, thereby effectively removing the a1 voltage regulation loop from the circuit. if the output current greater than 1 amp persists, the current limit loop forces a reduction of current to the load, which causes a corresponding drop in output voltage. as the output voltage drops, the current limit threshold also drops fractionally, result- ing in a decreasing output current as the output voltage de- creases, to the limit of less than 0.2 a at 1 v output. this fold-back effect reduces the power dissipation considerably during a short circuit condition, thus making the power supply far more forgiving in terms of the thermal design requirements. small heat sinking on the power mosfet can be tolerated. the op295s rail-to-rail swing exacts higher gate drive to the power mosfet, providing a fuller enhancement to the transistor. the regulator exhibits 0.2 v dropout at 500 ma of load current. at 1 amp output, the dropout voltage is typically 5.6 volts. 124k 1% 124k 1% ref43 4 6 2 0.01 m f 100k 5% 210k 1% 205k 1% 45.3k 1% 45.3k 1% r sense 0.1 w 1/4w +5v v o i o (norm) = 0.5a i o (max) = 1a s d g irf9531 1n4148 2.500v a1 a2 1 3 2 8 7 4 5 6 6v 1/2 op295/ op495 1/2 op295/ op495 figure 12. low dropout, 500 ma voltage regulator with fold-back current limiting square wave oscillator the circuit in figure 13 is a square wave oscillator (note the positive feedback). the rail-to-rail swing of the op295/op495 helps maintain a constant oscillation frequency even if the sup- ply voltage varies considerably. consider a battery powered sys- tem where the voltages are not regulated and drop over time. the rail-to-rail swing ensures that the noninverting input sees the full v+/2, rather than only a fraction of it. the constant frequency comes from the fact that the 58.7 k w feedback sets up schmitt trigger threshold levels that are di- rectly proportional to the supply voltage, as are the rc charge voltage levels. as a result, the rc charge time, and therefore the frequency, remains constant independent of supply voltage. the slew rate of the amplifier limits the oscillation frequency to a maximum of about 800 hz at a +5 v supply. single supply differential speaker driver connected as a differential speaker driver, the op295/op495 can deliver a minimum of 10 ma to the load. with a 600 w load, the op295/op495 can swing close to 5 volts peak-to-peak across the load.
op295/op495 rev. b C11C 8 1 4 2 3 58.7k 100k c r v+ freq out f osc = < 350hz @ v+ = +5v 1 rc 100k 1/2 op295/ op495 figure 13. square wave oscillator has stable frequency regardless of supply changes v in 10k v+ 10k 100k 90.9k 20k 20k v+ speaker 2.2 m f 1/4 op295/ op495 1/4 op295/ op495 1/4 op295/ op495 figure 14. single supply differential speaker driver high accuracy, single-supply, low power comparator the op295/op495 makes an accurate open-loop comparator. with a single +5 v supply, the offset error is less than 300 m v. fig- ure 15 shows the op295/op495s response time when operating open-loop with 4 mv overdrive. it exhibits a 4 ms response time at the rising edge and a 1.5 ms response time at the falling edge. 10 100 0% 90 5ms 2v 1v output (5mv overdrive @ op295 input) input figure 15. open-loop comparator response time with 5 mv overdrive op295/op495 spice model macro-model * node assignments * noninverting input * inverting input * positive supply * negative supply * output * * .subckt op295 1 2 99 50 20 * * input stage * i1 99 4 2e-6 r1 1 6 5e3 r2 2 5 5e3 cin 1 2 2e-12 ios 1 2 0.5e-9 d1 5 3 dz d2 6 3 dz eos 7 6 poly (1) (31,39) 30e-6 0.024 q1 8 5 4 qp q2 974qp r3 8 50 25.8e3 r4 9 50 25.8e3 * * gain stage * r7 10 98 270e6 g1 98 10 poly (1) (9,8) C4.26712e-9 27.8e-6 eref 98 0 (39, 0) 1 r5 99 39 100e3 r6 39 50 100e3 * * common mode stage * ecm 30 98 poly(2) (1,39) (2,39) 0 0.5 0.5 r12 30 31 1e6 r13 31 98 100 * * output stage * i2 18 50 1.59e-6 v2 99 12 dc 2.2763 q4 10 14 50 qna 1.0 r11 14 50 33 m3 15 10 13 13 mn l=9e-6 w =102e-6 ad=15e-10 ad=15e-10 m4 13 10 50 50 mn l=9e-6 w=50e-6 ad=75e-11 as=75e-11 d8 10 22 dx v3 22 50 dc 6 m2 20 10 14 14 mn l=9e-6 w=2000e-6 ad=30e-9 as=30e-9 q5 17 17 99 qpa 1.0 q6 18 17 99 qpa 4.0 r8 18 99 2.2e6 q7 18 19 99 qpa 1.0 r9 99 19 8 c2 18 99 20e-12 m6 15 12 17 99 mp l=9e-6 w=27e-6 ad=405e-12 as=405e-12 m1 20 18 19 99 mp l=9e-6 w=2000e-6 ad=30e-9 as=30e-9 d4 21 18 dx v4 99 21 dc 6 r10 10 11 6e3 c3 11 20 50e-12 .model qna npn (is=1.19e-16 bf=253 nf=0.99 vaf=193 ikf=2.76e-3 + ise=2.57e-13 ne=5 br=0.4 nr=0.988 var=15 ikr=1.465e-4 + isc=6.9e-16 nc=0.99 rb=2.0e3 irb=7.73e-6 rbm=132.8 re=4 rc=209 + cje=2.1e-13 vje=0.573 mje=0.364 fc=0.5 cjc=1.64e-13 vjc=0.534 mjc=0.5 + cjs=1.37e-12 vjs=0.59 mjs=0.5 tf=0.43e-9 ptf=30) .model qpa pnp (is=5.21e-17 bf=131 nf=0.99 vaf=62 ikf=8.35e-4 + ise=1.09e-14 ne=2.61 br=0.5 nr=0.984 var=15 ikr=3.96e-5 + isc=7.58e-16 nc=0.985 rb=1.52e3 irb=1.67e-5 rbm=368.5 re=6.31 rc=354.4 + cje=1.1e-13 vje=0.745 mje=0.33 fc=0.5 cjc=2.37e-13 vjc=0.762 mjc=0.4 + cjs =7.11e-13 vjs=0.45 mjs=0.412 tf=1.0e-9 ptf=30) .model mn nmos (level=3 vto=1.3 rs=0.3 rd=0.3 + tox=8.5e-8 ld=1.48e-6 nsub=1.53e16 uo=650 delta=10 vmax=2e5 + xj=1.75e-6 kappa=0.8 eta=0.066 theta=0.01 tpg=1 cj=2.9e-4 pb=0.837 + mj=0.407 cjsw=0.5e-9 mjsw=0.33) .model mp pmos (level=3 vto=C1.1 rs=0.7 rd=0.7 + tox=9.5e-8 ld=1.4 e-6 nsub=2.4e15 u o=650 delta=5.6 vmax=1e5 + xj=1.75e-6 kappa=1.7 eta=0.71 theta=5.9e-3 tpg=C1 cj=1.55e-4 pb=0.56 + mj=0.442 cjsw=0.4e-9 mjsw=0.33) .model dx d(is=1e-15) .model dz d (is=1e-15, bv=7) .model qp pnp (bf=125) .ends
rev. b C12C op295/op495 c1806aC10C7/95 printed in u.s.a. outline dimensions dimensions shown in inches and (mm) 8 lead plastic dip (p suffix) 0.160 (4.06) 0.115 (2.93) 0.130 (3.30) min 0.210 (5.33) max 0.015 (0.381) typ 0.430 (10.92) 0.348 (8.84) 0.280 (7.11) 0.240 (6.10) 4 5 8 1 0.070 (1.77) 0.045 (1.15) 0.022 (0.558) 0.014 (0.356) 0.325 (8.25) 0.300 (7.62) 0 - 15 0.100 (2.54) bsc 0.015 (0.381) 0.008 (0.204) seating plane 0.195 (4.95) 0.115 (2.93) 14-lead plastic dip (p suffix) pin 1 0.280 (7.11) 0.240 (6.10) 7 8 14 1 0.210 (5.33) max 0.160 (4.06) 0.115 (2.92) 0.795 (20.19) 0.725 (18.42) 0.022 (0.558) 0.014 (0.36) 0.100 (2.54) bsc 0.070 (1.77) 0.045 (1.15) seating plane 0.130 (3.30) min 0.015 (0.381) min 0.325 (8.25) 0.300 (7.62) 0.015 (0.38) 0.008 (0.20) 15 0 8-lead narrow-body so (s suffix) 0.0098 (0.25) 0.0075 (0.19) 0.0500 (1.27) 0.0160 (0.41) 8 0 0.0196 (0.50) 0.0099 (0.25) x 45 pin 1 0.1574 (4.00) 0.1497 (3.80) 0.2440 (6.20) 0.2284 (5.80) 4 5 1 8 0.0192 (0.49) 0.0138 (0.35) 0.0500 (1.27) bsc 0.0688 (1.75) 0.0532 (1.35) 0.0098 (0.25) 0.0040 (0.10) 0.1968 (5.00) 0.1890 (4.80) 16-lead wide-body so (s suffix) pin 1 0.2992 (7.60) 0.2914 (7.40) 0.4193 (10.65) 0.3937 (10.00) 1 16 9 8 0.0192 (0.49) 0.0138 (0.35) 0.0500 (1.27) bsc 0.1043 (2.65) 0.0926 (2.35) 0.4133 (10.50) 0.3977 (10.10) 0.0118 (0.30) 0.0040 (0.10) 0.0125 (0.32) 0.0091 (0.23) 0.0500 (1.27) 0.0157 (0.40) 8 0 0.0291 (0.74) 0.0098 (0.25) x 45

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